Reconfigurable, bi-directional, multi-band front end for a hybrid beamforming transceiver

ABSTRACT

Designs and techniques to enhance power-efficiency and incorporate new features in millimeter-wave MIMO transceivers are described. A new mechanism for built-in dual-band, per-element self-interference cancellation (SIC) is introduced to enable multi-antenna frequency-division duplex (FDD) and full-duplex (FD) operation. Additionally, several innovative circuit concepts are introduced, including low-loss wideband antenna interface design, dual-band power combining PA, dual-band RF-SIC design, and bi-directional MIMO signal

RELATED APPLICATIONS

This application is a continuation of U.S. patent application Ser. No.16/780,535 filed Feb. 3, 2020, which claims the benefit of U.S.Provisional Patent Application No. 62/918,505, filed Feb. 1, 2019, andwhich is a continuation-in-part of U.S. patent application Ser. No.16/677,072, filed Nov. 7, 2019. All applications in the chain ofpriority are incorporated herein in their entireties.

GOVERNMENT RIGHTS

This invention was made with U.S. government support under ECCS1343324awarded by the National Science Foundation. The government has certainrights in the invention.

BACKGROUND OF THE INVENTION

Integrated phased-arrays in silicon technology have emerged recently formillimeter-wave 5G and Wi-Fi communication. Typical phased-arrays form asingle steerable beam of electromagnetic radiation, thereby enablinghighly directional communication supporting a single data-stream. Futurewireless systems beyond 5G will likely usemultiple-input-multiple-output (MIMO) techniques to transfer multiplestreams of data simultaneously and thus can further improve data rate,device density and network capacity. In current phased-arrays, thesignal processing required to form and steer beams is predominantly doneusing radio frequency (RF)-domain circuits. However, the signalprocessing required to support MIMO is complex and, as such,phased-arrays do not efficiently support MIMO. On the other hand,digital beamformers (DBF) can enable high-complexity signal processingrequired for MIMO, but their power consumption is prohibitive. Hybridbeamforming (HBF) transceivers break this deadlock by partitioning therequisite signal processing between the RF and digital domains.Currently, all groups in academia and industry focus on“partially-connected” HBF's. “Fully-connected” HBF's are known to bemore efficient and offer superior performance, but they suffer from highimplementation complexity.

Millimeter-wave communication based on beamforming andmulti-input-multi-output (MIMO) techniques is expected to be arevolutionary new element in fifth-generation (5G) and beyond-5Gwireless networks. Many RF beamforming systems (i.e., phased arrays)have been demonstrated for a single-band within which they support asingle communication stream. Multi-element, multi-stream mm-wave MIMOover a single band are known in the art. The so-called fully-connectedhybrid beamforming (FC-HBF) architecture was introduced and shown toachieve superior energy-efficiency than partially connected HBF (PC-HBF)beamforming receiver, where a conventional phased array is used for eachstream.

FC-HBF architecture has been extended to a reconfigurable multi-band,MIMO/beamforming receiver, in which image-rejection heterodynedownconversion was used in conjunction with Cartesian-combining basedbeamforming to demonstrate multi-standard MIMO/beamforming in or both oftwo widely separated millimeter-wave bands the (for example, 27.5-28.35and/or 37-40 GHz) for potential 5G or beyond-5G mm-wave deployment. Inaddition to supporting MIMO in each band, the dual-band FC-HBF receiverarchitecture also supports an efficient inter-band multi-antenna carrieraggregation (CA) mode where each aggregated carrier could access thefull antenna aperture, and hence achieved full beamforming gain.

HBF receiver designs are known that are not of the FC kind. Moreover, incontrast to the aforementioned HBF approaches, multi-stream receiverdesigns that perform intermediate frequency (IF) beamforming in singleor multiple bands are known. An adaptive beamforming technique thatenables optimal minimum-mean-square-error (MMSE) beam/null steeringwithin a hardware constrained RF/hybrid beamformer is also known.

SUMMARY OF THE INVENTION

The invention described herein comprises a circuit architecture for abi-directional transmit/receive (T/R) module that can be used as afront-end interface between the radio electronics and an antenna array.It has the following salient features: (1) It can be used in manydifferent types of beamformers, including partially-connected HBF's,fully-connected HBFs and DBF's; (2) The circuit can be designed to havea single contiguous frequency response or a dual-band frequency responsecovering disparate frequency bands (e.g., 28/37 GHz). The dual-bandembodiment can support carrier aggregation (CA) where the circuit cantransmit or receive two signals concurrently in two bands of operation,thereby offering a mechanism to increase throughput; (3) Bothsingle-band and dual-band embodiments can be reconfigured for manydifferent types of system scenarios including time-division duplex MIMO,frequency-division duplex MIMO and multi-antenna CA; (4) The circuitfeatures a built-in mechanism to perform interference cancellation ofnarrowband interference in the front-end of the receive path of abeamforming transceiver. Different forms of interference can besuppressed using this mechanism, including “self-interferencecancellation (SIC)”; (5) The SIC mechanism extends the applicability ofthe proposed circuit to “simultaneous transmit-receive operation (STAR)”operation. Two forms of STAR can be supported. A STAR-Frequency Duplexed(STAR-FDD) mode, where the transmitter and receiver tuned to differentfrequency bands, and a STAR-Full Duplex mode (STAR-FUD) mode, where thetransmitter and receiver tuned to the same frequency channel in the sameband; and (6) In both STAR-FDD and STAR-FUD modes, signal leakage fromthe transmitter can corrupt the signals passing through the receiver anddestroy the fidelity of the received signal. This SIC inherently offersa mechanism to cancel such self-interference that protects the receivesignal from being corrupted.

Disclosed herein is a 28/39 GHz front-end applicable to beyond-5Gwireless networks. The invention exhibits three key features. First, afully-connected (FC) transmitter architecture is introduced for hybridbeamforming (HBF). It is shown that the power efficiency of FC-HBF issuperior to the conventional partially-connected (PC) HBF for a givenmodulation and antenna geometry. Second, a compact/low-cost circuitconcept is introduced that supports bi-directional T/R operationconcurrently at 28 and 37/39 GHz, thereby facilitating multi-antennacarrier-aggregation (CA) or MIMO time division duplexing (TDD) with highantenna count. Third, a built-in mechanism for dual-band, per antenna,self-interference cancellation (SIC) is introduced, thanks to the FC-HBFarchitecture. The front-end is applicable to FDD or full-duplex (FD)multi-antenna systems; such SIC is not available in PC-HBF's. Also, thefront-end is directly applicable to dual-band digital beamformers (DBF).

Also disclosed herein are two innovative system concepts for beyond-5Gmulti-antenna systems. First, the increase in peak-to-average powerratio (PAPR) in a FC-HBF (see View (B) of FIG. 1) transmitter comparedto PC-HBF see View (A) of FIG. 1) is identified for the first time. Itis shown that digital beamformers (DBF) have the same PAPRconsiderations as FC-HBF for an equivalent number of streams. To achieveidentical spectral efficiency, FC-HBF consumes significantly lower powerthan PC-HBF when any power amplifier topology that has a better back-offefficiency characteristic than Class-A is used.

Second, a new architecture which is conceived to directly enable forsimultaneous transmit and receive (STAR) beamforming for multi-antennafrequency-division-duplexing (FDD) or full-duplex (FD) communication isintroduced. The proposed FC-HBF structure can be reconfigured to enableself-interference cancellation (SIC) on a per-element basis without anyhardware overhead. Such SIC is not possible in a PC-HBF.

A compact circuit implementation is also introduced herein. The circuitarchitecture can support bidirectional transmit or receive operation at28 GHz, 37/39 GHz, or concurrently in both bands. Reconfiguration acrossmultiple bands helps avoid the use of a dedicated beamforming module foreach band, thereby reducing overall system complexity, area, and cost.

Three main configuration modes and ten sub-modes are available. Mode I(see View (A) of FIG. 2) and Mode II (see View (B) of FIG. 2) are timedivision-duplexed (TDD) FC-HBF transmit and receive modes, respectively,and can be configured for multi-stream MIMO operation in each band (insub-modes A and B) or inter-band CA (in sub mode C). Mode III (see View(C) of FIG. 2) enables FDD/FD, where half of the array is configured asTX and another half as RX with built-in SIC.

A bidirectional two-stream front-end prototype has been designed.Passive structures are extensively reused in the TX and RX modes toreduce die area.

This invention can serve as the front-end circuit supporting a singleantenna and one or more streams. In a MIMO transceiver, it can bereplicated for each antenna in an antenna array. The front-end featuresseveral new circuit techniques including: (1) a multi-band, low-lossLNA-PA-antenna interface network, (2) power-combining Class-B PA withdual-band second harmonic shorting network, (3) self-neutralizedbidirectional programmable-gain amplifier (PGA), (4) reconfigurablecombiner/splitter to support incorporation into FC-HBF transceivers, and(5) dual-band RF self-interference (SI) canceller.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 shows partially and fully connected hybrid beamformingtransceivers.

FIG. 2 shows three modes of operation for the fully connected hybridbeamforming transceiver showing fully connected transmit, fullyconnected receive and full-duplex (with SIC) modes.

FIG. 3 shows various characteristics of a partially-connected versus afully-connected HBF TX.

FIG. 4 shows a 28/37 GHz bidirectional-transmit-receive fully-connectedchannel.

FIG. 5 shows a dual-band antenna interface switch in view (1) and asecond harmonic short design in view (2).

FIG. 6 shows a transmitter path complex-weight and EVM in view (A) andFC versus PC DC power consumption (P_(DC)) in two stream FC/HBF andmultistream FC-HBF/DBF in view (B).

FIG. 7 shows FDD/FD mode dual-band per-element built-in SIC in FC-HBFTRX in view (A) and SIC versus phase and power of self-interferer, FDDand FD mode cancellation in view (B).

FIG. 8, view (A) is a schematic diagram of a shared-antenna STAR system.View (B) is a schematic diagram of a separate-antenna STAR system.

FIG. 9, view (A) is a schematic diagram of a multi-antenna STAR systemwith DBF and analog domain SIC. View (B) is a schematic of amulti-antenna STAR system with HBF featuring per-element SIC. View (C)shows antenna coupling amplitude and phase variation resulting singletap estimation air across frequency for two different antenna couplingscenarios.

FIG. 10, view (a) is a circuit diagram of a coupled resonator load. View(b) is a graph showing the ratio of the peak amplitudes in a dual-bandZ₁₁ and Z₂₁-based coupled resonator load. View (c) is a graph showingthe Z₁₁ and Z₂₁ frequency response for two operating regions.

FIG. 11, view (a) circuit diagram for a single band second harmonictrap. View (b) is a circuit diagram for a dual-band second harmonic trapnetwork used in the design shown in FIG. 4. View (c) is a graph showinga simulated door-Ben second harmonic short response.

FIG. 12 is a schematic circuit diagram showing a bidirectionalprogrammable gain stage. A forward path is shown in red in a backwardpath is shown in blue. The feedback path from both the output ports toone of the input ports is shown in green.

FIG. 13 shows an LNA/PA and stream #1/#2 interface splitter combinernetwork in two-stream FC/HBF TX mode in view (a), in two-stream FC/HBF'sRX mode in view (b) and in FDD or full-duplex beamforming mode with SICin view (c). View (d) is a graph showing the magnitude response of thecoupled-resonator load in the forward signal path (Z₂₁) and in thecancellation path (Z₂₂) showing the capability of door-band SIC.

DETAILED DESCRIPTION

Described herein is a 28/39 GHz front-end applicable to beyond-5Gwireless networks. Note that, while the invention is being explained interms of an implementation using 28/39 GHz, it should be realized by oneof skill in the art that the scope of the invention is intended toinclude any two mm-wave bands.

The front-end described herein features three novel aspects. First, afully-connected (FC) transmitter architecture is introduced for hybridbeamforming (HBF). The power efficiency of FC-HBF is superior to theconventional partially-connected (PC) HBF for a given modulation andantenna geometry. Second, a compact/low-cost circuit concept isintroduced that supports bi-directional T/R operation concurrently attwo mm-wave bands, thereby facilitating multi-antennacarrier-aggregation (CA) or MIMO TDD with high antenna count. Third, abuilt-in mechanism for dual-band, per antenna, self-interferencecancellation (SIC) is introduced, made possible by the FC-HBFarchitecture. The front-end is applicable to FDD or full-duplex (FD)multi-antenna systems. The described SIC is not available in PC-HBFs.Also, the front-end is directly applicable to dual-band digitalbeamformers (DBF).

Compared to a PC-HBF, the FC-HBF has superior energy and spectralefficiencies for a given antenna array geometry. View (B) of FIG. 1shows the proposed bi-directional FC T/R architecture and, in Views(A-C) of FIG. 2, how it can be reconfigured between 3 modes and 10sub-modes.

Modes I and II shown in Views (A-B) of FIG. 2 are FC-TX and FC-RX TDDmodes, respectively, and can be used for MIMO (in each band) or CA. ModeIII, shown in Views (C) of FIG. 2, applies to FDD/FDD, where a completeMIMO transceiver (also see FIG. 7) that comprises several of theproposed front-ends is partitioned into transmit (TX) and receive (RX)parts. In the TX part, the complex-weights in one of the streams areleft unused. In the RX part, one of the complex-weights is configuredfor reception, while the other complex-weight (which is available in theFC architecture) is used to inject an independently weighted SIC signalfor that element. Thus. SIC can relax the dynamic range requirement ofthe following RX circuits. This SIC technique provides an elegant way togenerate an independently weighted RF domain SIC signal from a singleupconverted TX signal for each antenna in an FDD or FD beamformer.

The pros and cons of FC-TX versus PC-TX are described next and aredisplayed graphically in FIG. 3. In a multiantenna system, the EIRPscales quadratically with the number of antennas. Because each stream(N_(S)=# of streams) accesses all N antennas in FC-TX and N/N_(S)antennas in PC-TX, FC-TX offers superior spatial combining.Consequently, each PA in FC-TX requires N_(S) times lower average outputpower to emit the same EIRP per stream as PC-TX. However, in an FC-TX,because independent signals from different streams are combined beforethe PA, the PAPR requirement of each PA is higher than themodulation-dependent PAPR of each stream. It can be shown that theeffective PAPR that must be handled by each PA is N_(S) times higher inFC-TX than in PC-TX. Therefore, the peak power specification of each PAin PCTX and FC-TX are equal. Based on these considerations, it isevident that for any PA that has lower DC consumption at back-off thanat peak output power (i.e., non-Class-A), the total power consumption ofFCTX is lower than PC-TX. Also, the margin of superiority of powerefficiency of FC-TX depends on the PA's back-off efficiency and improveswith N_(S). As shown in FIG. 3, with ideal Class-8 PA's, FC-TX has0.64×/0.6× the power consumption of PC-TX for QPSK/1024-QAM for twostreams while achieving identical spectral efficiency.

FIG. 4 shows a bi-directional, dual-band FC-HBF comprising severalconstituent circuit blocks designed for dual-band operation: a TX pathPA (blue), an RX path LNA (red), a shared antenna interface network(AIN), a splitter-combiner structure, independent bi-directionalcomplex-weight circuits for each MIMO stream and auxiliary on-chip testcircuits. It should be noted that the same structure can be designed tooperate in a single contiguous band.

The AIN, shown as (1) in FIG. 5, acts as a two-way transformerpower-combiner in TX mode and as a matching network to a CG LNA in RXmode. Switches S1-S2 and switches S3-S4 can be turned on or off toactivate a high-power (HP) TX mode, a low-power (LP) TX mode or the RXmode. Switches S1, S2, S3 and S4 may, in some embodiments, beimplemented as thin-gate or thick-gate MOS switches.

Each PA slice comprises three driver stages followed by an output stagewhich drives the primary of the power-combiner. In preferredembodiments, a Class B output stage may be used, while in otherembodiments, an output stage of any class may be used.

The Class-B stages employ a dual-band 2^(nd) harmonic shorting network,shown as (2) in FIG. 5, to improve efficiency and P_(SAT). In thisnetwork, the impedance Z_(X) is equivalent to an inductor havingdifferent values at the 2nd harmonic of the two frequency bands (56, 74GHz). Thus, the CM input impedance is equivalent to a series LC resonantnetwork that concurrently acts as a short at the two frequencies.

The signal path is designed so that additional T/R switches are notrequired over those used in the AIN. This includes the interface betweenthe LNA-PA and the combiner-splitter, and also the bi-directional gainstages in the PGA and the complex-weights, designed as back-to-backtransconductors, which are powered on or off to activate the TX or RXmodes.

The auxiliary circuits shown in FIG. 4 were included for testingpurposes in the present embodiment, but similar circuits can be used toexpand this front-end to a complete beamformer.

Circuit design choices have been made to achieve maximal compactness,thereby reducing chip area and hence the cost of the MIMO transceiver.Moreover, designing for compactness is advantageous since it reducesinterconnect lengths when such front-ends are combined into a largerMIMO transceiver. In particular, in one embodiment, passive structureswere: (1) shared between TX and RX; and (2) custom designed to minimizeslice height.

Simultaneous Transmit and Receive (STAR) Beamforming

STAR operation in separate transmit/receive frequency bands isequivalent to frequency division duplex (FDD), while STAR operation inthe same frequency band is also called full-duplex (FD), and results indoubling in throughput (a theoretical maximum) compared to a timedivision duplexed (TDD) system. In both FDD and FD, the key challenge inSTAR communication is the self-interference (SI) due to leakage of thestrong transmit signal into the path of the weak received signal,causing severe corruption through interference and non-linearity. In thecase of FDD, the leakage can be partially attenuated by filtering in thefront-end diplexer. However, this mechanism is not available in the FDcase, and therefore, signal cancellation of the transmit signal leakageis the only viable option.

There are two variants of STAR systems: shared-antenna STAR, shown inView (A) of FIG. 8, where each antenna element is shared betweentransmit and receive paths, and separate-antenna STAR, shown in View (B)of FIG. 8, where transmit and receive paths use completely separateantennas. While the shared-antenna approach has gained interest in sub-6GHz STAR communication, the separate-antenna approach is advantageous atmm-wave multi-antenna beamforming system for the following reasons: (1)The separate-antenna approach avoids the use of a circulator. On-chipimplementations of circulators are possible, but they are lossy, havelimited linearity and bandwidth, and achieve inadequate transmit-receiveisolation. Furthermore, they occupy large die area and are difficult tointegrate into multi-antenna transceivers with large numbers of elementsin a cost-effective manner; (2) More importantly, due to small elementspacing at mm-wave, adjacent elements experience significant coupling.Hence, inter-element self-interference (SI) between nearby elementswould be severe in the shared-antenna approach. In a circulator-basedSTAR beamforming system, although the circulator in each elementisolates the receiver from SI from its own transmit signal, SI fromadjacent elements pass through the circulator with little attenuation.However, in a separate-antenna approach, SI due to antenna coupling canbe greatly reduced by increasing the physical spacing between thetransmit and receive antenna arrays.

A single separate-antenna STAR transceiver, shown in View (B) OF FIG. 8,can be extended to a multi-antenna transceiver as shown in View (A) ofFIG. 9. The transmit path has a DAC, upconverter and PA in each elementalong with digital precoding, while the receive path has LNA,downconverter and ADC in each element along with digital beamforming.However, the conventional method of performing RF SIC by feeding aweighted PA signal to the canceller path is impractical in amulti-antenna system because feedback from each TX antenna to every RXantenna would be required. Therefore, as shown in View (A) of FIG. 7, anRF domain cancellation signal must be generated separately for eachelement in the RX array by using an upconverter and DAC per element toperform per element SIC. The back end digital processing can furtherreduce the SI by optimizing the TX leakage in the TX array and bycancelling the residue SI in the digital domain in the RX array.

View (B) of FIG. 9, shows how the aforementioned SIC approach can beimplemented in the proposed bi-directional FC-HBF (see View (C) OF FIG.2). Here, a reduced number of bidirectional frequency translation chains(i.e., streams) interface to a large number of antennas throughper-element bi-directional RF-domain complex weights. Two separateFC-HBF transceivers are used in the FD transceiver, one each for TX andRX. In the TX FC-HBF, one stream is configured for transmit. In the RXpart, one stream is configured for receive, while the other stream isrepurposed to upconvert a copy of the baseband transmit signal. Theupconverted transmit signal copy can be independently complex weightedin each element to cancel the incoming transmit signal leakage, therebyperforming a per-element SIC. Thanks to its built-in SIC mechanism, theFC-HBF transceiver offers an efficient basis for STAR beamformingwithout the need for additional cancellation circuitry. It is importantto note that a similar SIC mechanism is not available in the PC-HBF.

The SI in a multi-antenna STAR system can occur in two ways: (1) SIthrough antenna coupling from the transmit to the receive antennas; and(2) SI due to a nearby reflection of the transmitted signal that leaksinto the receiver through the receive antenna array. The first kind ofSI has small group delay, where the second kind may have small or largegroup delay depending on the distance of the reflector. The SI withlower group delay is expected to have higher strength due to havinglower path loss. While the DBF-based STAR system in View (A) of FIG. 9can cancel SI with both small and large group delays, the FC-HBF-basedSTAR system shown in View (B) of FIG. 9 can only cancel SI with smallgroup delay as only a single tap-independent cancellation can beperformed in each element.

View (C) of FIG. 9 shows the measured coupling between patch elements(same polarization) for two different multi-antenna configurations. Overa 500 MHz signal bandwidth, the coupling is seen to have fairly flatamplitude and phase responses, which can be estimated by a single tap(i.e., αe^(jθ)) with better than −40 dB of estimation error. However, inthe FC-HBF, large group-delay SI can be canceled by directing aspatial-null towards the reflection paths in both the transmit andreceive beamformer (at the cost of some degradation of the main beamgain in case of low element count arrays). Such SIC, based onnull-steering in the RX array cancels the SI only after beamforming; onthe other hand, the single-tap RF SIC cancels the SI right at LNAoutput. Note that, nulls in both the transmit and receive array patternscan be steered towards different leakage multipath components or can besteered towards the same path to achieve higher rejection.

FIG. 7, view(A) shows a conceptual FDD/FD beamforming node including TX(blue). RX (red) and per-antenna SIC (green). The RX signal passesthrough Z₂₁ of the coupled-resonator. The SIC signal current is injectedinto the Z₂₂ port of the coupled-resonator using a complex-weight cell(5 b I/Q) followed by fine gain control PGA (5 b, no sign) and finephase control (<1°). Z₂₁ and Z₂₂ arm designed to have similar frequencyresponses at the resonant frequencies (placed at the bands of interest)to enable dual-band SIC. To characterize SIC performance, the concept ofFIG. 7 was simulated using external equipment to generate the RX signal,the SI and a reference version of the SI leakage. SIC greater than 36 dBwas measured over 360° SI signal phase variation and over 36 dB range ofSI power, as shown in FIG. 7, View (B).

Next, FDD operation is characterized using a desired 37 GHz RX tone anda two-tone interferer near 28 GHz. SIC weight settings were set once formaximum cancellation at band center. The two-tone spacing was swept tocharacterize SIC over BW. FIG. 7 shows that 28d8 SIC was achieved for0.4 GHz RF BW. FD mode operation is then demonstrated at 28 GHz using adesired RX tone and a modulated SI; 26 dB SIC is achieved over 500 MHzRF BW of the interferer.

28/39 GHZ FC-HBF Circuit Design

A reconfigurable bidirectional multi-band (28/39 GHz) FC-HBF transceiverfront-end has been designed in a 65-nm CMOS process. A detailedschematic of the front-end is shown in FIG. 4 along with on-chip testcircuitry. The prototype constitutes a TX path (highlighted in blue inFIG. 4) dual-band two-way power combining power amplifier (PA) and an RXpath (highlighted in red in FIG. 4) low-noise amplifier (LNA) thatinterfaces the antenna port using a shared (shared blocks are shown inblack in FIG. 4) multi-band antenna interface network (AIN). Opposite tothe antenna port, LNA and PA are interfaced with two bidirectional T/Rstreams, where each stream consists of a complex weight, a fine gaincontrol, and a fine phase control. All large area passive structures inthe TX and RX paths, except those inside LNA and the PA are reused forcompactness.

Moreover, all the signal path reconfigurations are done without usingany switch in the signal path. Two sets of bi-directional PGAs areavailable for two-stream complex weighting in hybrid MIMO/beamforming.The bi-directional front-end can be used as the core building block of amulti-mode two-stream FC-HBF transceiver of the type shown in FIG. 1 andFIG. 2. This can be done in one of two ways: (1) by having onequadrature hybrid per element per stream after the PGA's, followed byone parallel combiner for each stream; or (2) using a dual-bandCartesian combining approach. Also, it should be noted that, byexcluding the complex weights, the front-end can be used directly in aDBF (View (A) of FIG. 9).

In the prototype, different off-chip interfaces are used for the weightsin each stream; this is done solely for test purposes. In stream #1, theI- and Q-PGA's are connected via baluns to separate pads for standalonetesting. In stream #2, a quadrature hybrid is incorporated to implementfull complex weighting.

Multi-Band Antenna Interface Network (AIN)

In this section, the design of a compact, multi-band antenna interfacenetwork is discussed. Consider a common single-band solution where theantenna interface switch is implemented using a series quarterwavelength (λ/4) transmission line and a shunt switch. Even at mm-wavefrequencies, on-chip λ/4 transmission lines have large footprint andhigh insertion loss, thereby degrading TX path output power (P_(out))and RX path noise figure (NF). Partial solutions to this problem areproposed in single-band (28 GHz) phased arrays, where the TX path λ/4line is eliminated, thereby avoiding the output power penalty. However,several shortcomings remain. A λ/4 line is still required in the RX pathand has large footprint and high loss. High inductance in the RX pathaffects the LNA input matching bandwidth. In the RX mode of bothdesigns, the RX input experiences an LC-tuned OFF-state TX load thatpresents high impedance at the antenna port only over a narrowbandwidth. This adversely affects the RX input match bandwidth. Moreimportantly, in TX mode, the OFF-state RX-side switch in View (a) ofFIG. 9 loads the TX via the λ/4 line, causing reduced TX bandwidth andoutput power. This loss can be estimated as follows. For a transmissionline with characteristic impedance Z₀ and quarter wavelength atfrequency f_(C) terminated in a switch with resistance RON, the inputimpedance can be written as:

${Z_{off}(f)} = {{Z_{0}\frac{R_{ON} + {jZ_{0}{\tan\left( {\frac{\pi}{2} \times \frac{f}{f_{C}}} \right)}}}{Z_{0} + {jR_{ON}{\tan\left( {\frac{\pi}{2} \times \frac{f}{f_{C}}} \right)}}}}\overset{R_{ON}\rightarrow 0}{\rightarrow}{{jZ}_{0}\left( {\frac{\pi}{2} \times \frac{f}{f_{C}}} \right)}}$

It is seen that the impedance looking into the RX is high at f_(C), buthas low reactive values as the operating frequency deviates from f_(C).If a PA is designed for the optimal output impedance of the antenna Z₀at f_(C), it can be shown that the PA's maximum output power at afrequency f in the presence of OFF-state RX side impedance is given by:

$\frac{P_{out}(f)}{P_{out}\left( f_{c} \right)} = {\left( {1 - \frac{Z_{0}^{2}}{Z_{OFF}^{2}(f)}} \right)^{1/2} = \left( {1 - {\cot^{2}\left( {\frac{\pi}{2} \times \frac{f}{f_{C}}} \right)}} \right)^{1/2}}$

Thus, if a single-band design is used in the dual-band design at hand,the OFF-state RX-side impedance can cause as high as ˜3 dB of loss atf=37 GHz for f_(C)=28 GHz. Simulation conducted with the designed powercombining PA shows similar output power degradation when a similarOFF-state switch is connected to the output of the PA.

View (1) of FIG. 5 show the various configuration modes of the proposedmulti-band AIN, where an LNA and a two-way power combining PA areinterfaced to an antenna port using a single passive network and threeMOS switches S₁,

TX Mode: In the TX mode, LNA input switch S₁ is in the ON-state. The AINitself serves as the two-way power combining network. In the high power(HP) TX mode switches S₃, and S₄ are both left open and both PA slicesturned ON. In the low power (LP) TX mode, only S₃ is open with one PAslice ON. This AIN design overcomes the bandwidth limitations ofconventional front-end switches. It can be shown that the non-zeroR_(ON) of the switch S₁ results in a loss of 20 log (1+R^(ON)/50) dB,which is only ˜0.3 dB for R_(ON)=2Ω. Similar loss is estimated insimulation. Note that, although the LNA is turned OFF in the TX mode byturning OFF all the LNA biases, a non-negligible feedback signal stillflows through the LNA from the PA's output to its input (see FIG. 4),which degrades PA stability. To further reduce the feedback signalstrength, the switch S₂ is turned ON (see FIG. 3a ), which significantlyreduces the gain of the first LNA stage.

RX Mode: In the RX mode, LNA input switch S₁ is turned OFF, S₃ and S₄are both turned ON, and both the PA slices OFF. Therefore, the RX pathconsists of a p-matching network with series inductance from AIN, theLNA input capacitance and the antenna port capacitance. It can be shownthat maximum series inductance L_(AIN) that can be used to match an LNAinput impedance R_(i,LNA) to antenna port input impedance of R_(i,ANT)at frequency f is the following:

max(L _(AIN))=L _(m)=√{square root over (R _(i,LNA) R _(i,ANT1))}/(2πf)

In the proposed design, turning ON switches S₃ and S₄ is especiallyadvantageous. This is because, in RX configuration, they help reduce theseries inductance by a factor of (1−k²) compared to when both switchesare OFF. Therefore, by turning ON S₃ and S₄, the series inductance isreduced below Lm which makes the input matching feasible. Ag_(m)-boosted common-gate input stage is used in the LNA. R_(i,LNA) isdesigned to be lower than 50Ω to reduce the input transistor's noisecontribution. In simulation, the π-matching network degrades the LNA NFonly by 0.8/1 dB in 28/37 GHz bands.

Two-Way Power Combining Power Amplifier (PA)

PA Core: RF and mm-wave PAs in low-voltage CMOS technology achieve highoutput power by coherently combining the outputs of multiple PA units.In such power combining PA's, a subset of PA units can be turned OFF toimprove back-off efficiency at lower input power. Herein, two-way powercombining is implemented using a transformer-based power combiningnetwork. As explained above, the PA can be configured into HP or LPmodes. By using the LP configuration at lower input power, PA back-offefficiency can be significantly improved, which in turn enhances thesuperiority of FC-HBF or DBF architectures.

The output stage of each PA unit is biased in deep class-AB (aroundclass-B) to improve peak-efficiency as well as back-off efficiency overclass-A PA. Moreover, as the third harmonic current from the outputstage transistors can be substantially reduced by biasing them inclass-B, the output 1-dB compression point can also be pushed closer tothe peak output power, thereby achieving flatter amplitude-to-amplitude(AM-AM) characteristics. Additionally, the sweet spot biasing aroundclass-B also reduces amplitude-to-phase (AM-PM) distortion even withoutthe varactor-based gate capacitance nonlinearity compensation. However,class-B output stage suffers from low gain. Therefore, three driverstages with class-AB biasing are used in each PA unit to ensure highoverall PA gain. The transconductors in the output stage and driverstages #1 and #2 (see FIG. 4) use pseudo-differential NMOS pairs withcross-coupled C_(GD) neutralization for improved linearity at a scaledpower supply. The driver stage #3 in FIG. 4) employs pseudo differentialpair with cascode for better output to input isolation. All inter-stagematching networks use dual-band coupled-resonator loads, whose design isexplained below. Common-mode stability of the PA is greatly improved byleaving the secondary side center tap of the transformers open (see FIG.4). This restricts the common-mode feedback current to flow to theprevious stage by providing a very high common mode impedance.

Dual-Band Loads and Gain Equalization: The transimpedance (Z₂₁) of atransformer coupled-resonator can be used to realize wideband load andto realize a dual-band load. Z₂₁-based dual-band loads are extensivelyused in this design in the PA as well as in other parts of thefront-end. In addition to Z₂₁, the driving point impedance (Z₁₁) of atransformer coupled-resonator also has a dual-band characteristic thatis used in the driver stage #3 (see FIG. 4) of the PA. The equationbelow reveals that both the Z₁₁ and Z₂₁ of a symmetricallycoupled-resonator (L₁=L₂=L and C₁=C₂=C in View (a) of FIG. 10) haveidentical complex pole pairs, and therefore can both be used asconcurrent dual-band loads where the center frequencies of the two bandscoincide with the two pole frequencies this equation:

${{{Z_{11}(S)} = \frac{\omega_{0}^{2}{L\left( {S + \frac{\omega_{0}}{Q}} \right)}\left( {S^{2} + {\frac{\omega_{z}}{Q_{z}}S} + \omega_{z}^{2}} \right)}{\left( {S^{2} + {\frac{\omega_{p1}}{Q_{p1}}S} + \omega_{p1}^{2}} \right)\left( {S^{2} + {\frac{\omega_{p2}}{Q_{p2}}S} + \omega_{p2}^{2}} \right)}}{Z_{21}(S)}} = \frac{kL\omega_{p1}^{2}\omega_{p2}^{2} \times S}{\left( {S^{2} + {\frac{\omega_{p1}}{Q_{p1}}S} + \omega_{p1}^{2}} \right)\left( {S^{2} + {\frac{\omega_{p2}}{Q_{p2}}S} + \omega_{p2}^{2}} \right)}$where${\omega_{{p1},{p2}} = \frac{\omega_{0}}{\sqrt{1 \pm k}}},{\omega_{z} = \frac{\omega_{0}}{\sqrt{1 - k}}},{\omega_{0} = \frac{1}{\sqrt{LC}}}$and${Q_{{p1},{p2}} = {Q\sqrt{1 \pm k}}},\ {Q_{z} = {Q\frac{\sqrt{1 - k^{2}}}{1 + k^{2}}}},{Q = \frac{\omega_{0}L}{R_{S}}}$

In the context of dual-band design, Z₁₁ and Z₂₁-based loads have thefollowing advantages and disadvantages. (1) In Z₂₁-based design thedrive port and the load port are isolated. Hence, the parasiticcapacitance of the drive and load ports can be separately absorbed inthe two sides of the coupled resonator. Therefore, Z₂₁-based design cansupport higher parasitic capacitance while using an identicaltransformer to achieve similar peak gain; (2) The Z₂₁-based design withtwo transformer feed points at opposite sides of the transformer coilcan be adopted to realize a long and skinny layout for each stage. Onthe other hand, in a Z₁₁-based design, as the driver and the load areboth connected to the same port of the transformer, Z₁₁ loads can beadopted in a scenario where signal path takes a 90° turn (see driverstage #3 in FIG. 4); (3) In contrast to a Z₂₁-based design where thehigh-frequency resonance mode (at ω_(H)) always has worse gain than thelow frequency mode (at ω_(L)), Z₁₁-based design with asymmetricresonator can achieve higher gain either at ω_(H) or at ω_(L). For anasymmetric resonator with desired dual-band operation at ω_(H) and atω_(L), resonator's design parameters w₁(=1/√{square root over (L₁C₁)})and ω₂(=1/√{square root over (L₂C₂)} can be chosen according to thefollowing equations, where k should be less than or equal to (ω_(H)²−ω_(L) ²)/(ω_(H) ²+ω_(L) ²).

$\omega_{1}^{2} = {\frac{1}{L_{1}C_{1}} = {\frac{\left( {1 - k^{2}} \right)}{2}\left\lbrack {\omega_{H}^{2} + {\omega_{L}^{2} \mp \sqrt{\left( {\omega_{H}^{2} + \omega_{L}^{2}} \right)^{2} - \frac{4\omega_{H}^{2}\omega_{L}^{2}}{\left( {1 - k^{2}} \right)}}}} \right\rbrack}}$$\omega_{2}^{2} = {\frac{1}{L_{2}C_{2}} = {\frac{\left( {1 - k^{2}} \right)}{2}\left\lbrack {\omega_{H}^{2} + {\omega_{L}^{2} \pm \sqrt{\left( {\omega_{H}^{2} + \omega_{L}^{2}} \right)^{2} - \frac{4\omega_{H}^{2}\omega_{L}^{2}}{\left( {1 - k^{2}} \right)}}}} \right\rbrack}}$

Note that ω₁ and ω₂ have two solutions—one for ω₁>ω₂ and other for ω₁<ω₂(shown in View (b) of FIG. 10 in the two sides of the dotted line). Tofind the design space for achieving higher or lower amplitudes at ω_(H)or ω_(L), the Z₁₁ and Z₂₁ amplitude responses are simulated for anasymmetric resonator with L₁=L₂=200ph and Q₁=Q₂=20. The ratio of twopeak amplitudes at ω_(L) and ω_(H) is plotted versus k in View (b) ofFIG. 10 in dB scale for f_(L)=28 GHz and f_(H)=38 GHz. It can be seenthat the amplitude ratio in case of Z₂₁ is always greater than 0 dB,where in case of Z₁₁ it can be either greater or less than 0 dB. Twoexemplary amplitude responses for two operating regions of Z₁₁ are shownin View (c) of FIG. 10. Therefore, it can be concluded that Z₁₁-baseddesign can help equalizing the gain at high- and low-frequency bands ina dual-band design. The aforementioned property is utilized in the PAdesign to reduce the gain difference in the two operating bands byincorporating one Z₁₁-based driver with other Z₂₁-based gain stages.

Dual-Band Second Harmonic Short: PA output stage biased in deep class-ABor in class-B generates significant second harmonic currents underlarge-signal condition. The second harmonic current when flows throughthe load impedance creates significant second harmonic voltage at theoutput node that can degrade P_(sat), PAE and AM-PM distortion of the PAshow that the performance degradation can be overcome by placingharmonic traps (load network that provides the harmonic current a lowimpedance path to ground) at the PA output node. However, previoustechniques employ a single frequency second-harmonic trap. Thesetechniques are difficult to incorporate in wideband or multibanddesigns. A conventional harmonic trap design is shown in View (a) ofFIG. 11. To support 28 and 37 GHz operation, significant amount offrequency tuning is required, thereby degrading quality factor of thetrap across frequency. Moreover, conventional harmonic-trap networkscannot support concurrent dual-band second harmonic trapping.

A novel second-harmonic trap network is introduced next. It utilizes atransformer-coupled resonator to realize a dual-band short, as shown inView (b) of FIG. 11. In a differential PA design, second harmoniccurrent appears as common mode signal. The common-mode equivalentnetwork of the proposed design is shown in View (B) of FIG. 11, wherethe common-mode input impedance (Z_(X1)) of the network can becalculated as follows.

${Z_{X1}(S)} = {2{L\left( {1 - k^{2}} \right)} \times \frac{\left( {S^{2} + {\frac{\omega_{Z1}}{Q_{Z1}}S} + \omega_{Z1}^{2}} \right)\left( {S^{2} + {\frac{\omega_{Z2}}{Q_{Z2}}S} + \omega_{Z2}^{2}} \right)}{S\left( {S^{2} + {\frac{\omega_{0}}{Q_{0}}S} + \omega_{01}^{2}} \right)}}$where${\omega_{z1} = \frac{1}{\sqrt{4\left( {1 + k} \right)LC}}},\ {\omega_{z2} = \frac{1}{\sqrt{4\left( {1 - k} \right)LC}}}$and${Q_{{z\; 1},{z\; 2}} = {Q_{0}\sqrt{\left( {1 \pm k} \right)}}},\ {Q_{0} = {\omega_{0}{L/R}}},\ {\omega_{0} = {1/\sqrt{4LC}}}$

The equation above reveals that Z_(X1) has two zeroes that concurrentlyprovide low-impedance paths at two frequencies. The proposed network isequivalent to a series LC network where the inductance (Z_(X) in View(b) of FIG. 11) concurrently takes two different values at two resonantmodes of the transformer (even and odd mode). A simulated dual-bandsecond-harmonic trap response is shown in View (c) of FIG. 11 for adual-band 56/4 GHz short. The second harmonic short reduces thesecond-harmonic current in the PA's output stage by ˜3×.

Bidirectional Self-Neutralized Phase-Invariant PGA

Unlike conventional phased-array transceivers where separate PGAs havebeen used in the TX and RX paths, all PGAs in this prototype aredesigned to share passives in TX and RX configuration for compactness.Compact designs not only reduce area and cost, but also eliminate lossesdue to long interconnects. A straightforward way to realize abidirectional PGA is by using a single programmable transconductor inconjunction with signal-path switches to reverse the direction of signalflow. However, in such a design, the signal-path switches can causesignificant loss at mm-wave frequency while also degrading PGAlinearity.

Bidirectional PGA Design: To overcome the aforementioned problem, abi-directional PGA design is introduced that avoids signal-pathswitches, as shown in FIG. 12. Here, two back-to-back programmabletransconductors are used, one for each signal path direction. Toconfigure for forward (reverse) signal flow, the forward (reverse)programmable transconductor is turned ON, while the reverse (forward)transconductor is turned OFF. In addition to turning ON/OFF thetransconductor cells by controlling the tail-bias of the differentialpair, separate biasing network for the transconductors' gate-bias isalso used to enable either the forward or the backward path. Althoughthe proposed technique uses twice the number of active devices andincreases the capacitance at each node by the OFF-state capacitance ofeach path, it eliminates signal-path switch loss.

Neutralization Technique: A differential amplifier without cascodedevices experiences output-to-input feedback through the gate-to-draincapacitance (C_(GD)) of the input pair. An explicit cross-coupledcapacitance pair can be used to neutralize this feedback, therebyimproving differential-mode stability. Neutralization based on thisprinciple is implicitly available in the proposed bi-directional PGAtopology, because the C_(GD) of the OFF-state transconductor in thereverse/forward path cancels the feedback through the ON-statetransconductor in the forward/reverse path (see FIG. 12).Transconductors in the forward and reverse paths should be oppositelyconnected, as shown in FIG. 12 (the feedback path from one output toboth the input nodes are highlighted in green). It is important to notethat feedback capacitances from ON and OFF paths in the self-neutralizedbidirectional architecture matches well over PVT, thereby providingPVT-invariant differential-mode stability.

Common-mode stability of the proposed bi-directional PGA topology isimproved by using switches S_(C1)-S_(C4) at the center taps of thecoupled resonators that selectively connects to the power supply orleave it open to reduce common mode feedthrough. As shown in FIG. 13,switches S_(C1) and S_(C3) (S_(C2) and S_(C4)) are turned ON for theforward (reverse) configuration.

Bidirectional Per-Stream Complex-Weighting

The design of the per-stream, bi-directional complex weights isdescribed with reference to FIG. 4. A pair of bidirectional 5-bit(including sign bit) PGAs is used in each stream to realize Cartesiancomplex weights. In the TX path, inputs to the PGA pair can be generatedlocally by using quadrature hybrids (one per stream) or can be generatedglobally before splitting to other elements by extending the Cartesiancombining technique. The outputs of the PGA's in the complex weight arecombined in the TX path using a bidirectional current-mode-combiner.Similarly, in the RX path, the inputs of the PGA's in the complex weightare generated using a voltage-mode-splitter, and the outputs from thePGA pair can be combined either locally or globally. Additionally, finegain control (˜0.3 dB LSB) in each stream is achieved by using anadditional 5-bit (without sign) bi-directional PGA, and fine phasecontrol (˜0.5°) is achieved by using a 5-bit digitally switchablecapacitor in the coupled resonator load of the PGA, as shown in FIG. 4.Fine gain and phase control enable accurate complex-weighting andaccurate SIC in the STAR mode. Finally, to improve common modestability, switches S₅-S₈ in FIG. 4 is incorporated in two streams.

Splitter/Combiner and Dual-Band SIC

Both the bi-directional streams are interfaced with the PA and the LNAusing a single coupled resonator. One side (e.g., the primary side) ofthe coupled resonator is connected to two PA slices and the LNA, and thesecondary side is connected to both streams (see FIG. 3a ). Views (a-c)of FIG. 13 show how the interface can be configured in TX, RX and STARmodes without signal path switches. In the TX mode, LNA is turned OFFand both streams are configured to transmit, as shown in View(a) of FIG.13. Signal currents from the two PGAs from the two streams are combinedin the secondary side of the coupled resonator. The voltage developed inthe primary side splits to two PA slices via the voltage-mode splitter.In the RX mode, both PA slices are turned OFF and both streams areconfigured to receive, as shown in View (b) of FIG. 13. The signalcurrent from LNA stage #2 transconductor is fed into the primary side ofthe coupled resonator. The voltage developed on the secondary side issplit into two streams by the voltage-mode splitter by reusing the samepassive structure that is used for current combining in the TX mode.

In the STAR mode of operation, per-element SIC is required in the RXbeamformer. In this mode, both PA slices are turned OFF. One stream isconfigured as receive for the desired input, and the other stream isconfigured as transmit to perform SIC, as shown in View (c) of FIG. 13.The signal current from LNA's stage #2 transconductor is inserted in theprimary side of the coupled resonator, and the signal current from theSIC path stream is fed into the secondary side. After SIC is performedinside the coupled-resonator, the voltage developed in the secondaryside drives the receive stream. Therefore, in STAR mode, the forwardpath sees the trans-impedance Z₂₁ of the resonator as the load, whilethe SIC path sees the driving port impedance (Z₂₂) as the load (see View(c) of FIG. 13). As described above, both the driving port impedance andthe transimpedance of a coupled resonator have dual-band behavior. Thefollowing equation shows how the gain through both the paths can beequalized in both bands by selecting moderate coupling and high kQproduct. This is supported by the simulation shown in View (d) of FIG.13.

${{\frac{Z_{11}\left( {j\;\omega} \right)}{Z_{21}\left( {j\;\omega} \right)}}_{\omega_{{p\; 1},{p\; 2}}} \approx {\frac{\omega_{z}^{2} - \omega^{2} + {j\frac{\omega_{z}}{Q_{z}}\omega}}{k\omega_{p1}^{2}{\omega_{p2}^{2}/\omega_{0}^{2}}}}_{\omega_{{p\; 1},{p\; 2}}}} = \sqrt{{1 + \left( {\frac{1}{KQ} \times \frac{1 + k^{2}}{\sqrt{1 \pm k}}} \right)^{2}} \approx 1}$

Thus, both the receive path and the SIC path can achieve reasonable gainconcurrently in both bands, thereby enabling cancellation in either ofthe two bands. The forward and the cancellation path can be at twodifferent frequencies (in FDD mode) or can be in the same frequency band(in FD mode). It should be noted that the noise from SIC path candegrade the RX NF in FDD/FD configuration. However, since the SIC isperformed after two LNA stages in the RX chain, NF degradation isminimal (0.4/0.5 dB in 28/37 GHz band in simulation) in this design.

A new multi-antenna simultaneous transmit-receive system architecturehas been introduced herein that provides a way to cancelself-interference in the RF-domain on a per-element basis in an FC-HBFor DBF transceiver. Additionally, a compact circuit topology isintroduced that realizes dual-band bi-directional operation whileintroducing minimal loss from the TX-RX switching networks.

Numerous innovative circuit techniques are disclosed herein thatincludes dual-band antenna interface, dual-band second harmonic shortingnetwork, dual-band gain equalization and bi-directional self-neutralizedPGA. The front-end design can be incorporated directly in a digital orhybrid beamforming transceiver system. The front-end achievesstate-of-the-art performance when benchmarked against recent 28 GHzbeamformers, multi-band mm-wave PAs, and single antenna STAR system withRF-domain SIC.

The embodiments have been explained in terms of specific designs.However, as would be realized by one of skill in the art, variations ofthe exemplary designs which are still within the scope of the inventionas defined by the following claims are possible.

We claim:
 1. A bi-directional, transmit/receive module suitable for useas a front-end interface between radio electronics and an antenna arrayin a beamforming transceiver, comprising: an amplification modulecoupled to the antenna array; a splitter/combiner module coupled to theamplification module transmit; and a bi-directional complex-valuedweighting module, coupled to the splitter/combined module; wherein thetransmit/receive module can be configured to operate in one of threemodes; a first mode wherein the transmit/receive module operates as afully-connected hybrid beamforming transmitter; a second mode whereinthe transmit/receive module operates as a fully-connected hybridbeamforming receiver; and a third mode wherein the transmit/receivemodule operates as full duplex or frequency-division duplex hybridbeamforming transceiver.